1. Introduction
Nowadays, advanced wireless systems or phased-array radars have raised the requirement of multifunctional antennas to perform communication, detection, and other tasks [1]-[9]. Therefore, researchers are trying to combine multiband antennas in the same radiation aperture, because multiband shared-aperture antennas can cover a wide operating bandwidth and support the simultaneous working situation in different frequency bands [10]-[25].
However, there are still some existing problems hindering the application of dual-band antenna in phased-array systems. Firstly, the bandwidth of the dual-band antennas is relatively narrow. For a typical Ku/Ka phased-array radar system, the operating bandwidth for both frequency bands should be no less than 4 GHz, corresponding to a relative bandwidth of 25% [26]. In the dual-band antenna array, the low-frequency antennas are placed in the gap of high-frequency ones, which will strictly limit the volume or area of a single element, and make it difficult to extend the bandwidth. As a result, the relative bandwidth is less than 14% in many cases [27]. Secondly, the beam scanning range is relatively small, especially for millimeter wave antennas. Due to the compact layout of dual-band antenna array, the element spacing is usually larger than that of the traditional phased array, which will directly limit the beam coverage. Meanwhile, the relatively high profile of the low-frequency antenna often causes deterioration of radiating performance of high-frequency antenna. Thirdly, the phased-array radar systems require good assemblability and channel isolation, which is difficult to realize for traditional dual-band antennas. For advanced dual-band phased-array radar systems, every feeding port of the elements needs to be connected to TR modules, and antennas of two frequency bands are usually required to work simultaneously. Nevertheless, external baluns are usually necessary for traditional wide-band or wide-angle scanning antennas, which makes it hard to integrate dual-band antennas in phased-array radar systems. Besides, the channel isolation of dual-band antennas is usually poor due to the compact arrangement, which may cause deterioration of the radiating performance.
In this paper, a Ku/Ka dual-band shared-aperture phased-array antenna with wide bandwidth and wide beam coverage is proposed. The dual-band antennas are combined with Ku-band metallic banyan tree antenna (BTA) elements and Ka-band printed circuit board-based (PCB-based) slot antenna elements. The Ku elements are optimized to achieve the operating bandwidth of 4 GHz and the scanning coverage of \(\pm 60^\circ\), and two square slots on the tapered edge are designed to realize the low-profile characteristic. Meanwhile, the feeding pin of Ku elements is integrated in the coaxial cable without external baluns, making it easy and flexible to connect to TR modules. The Ka elements are optimized to achieve the operating bandwidth of 4 GHz. Thanks to the low-profile Ku elements, the beam-scanning coverage of Ka elements achieves \(\pm 30^\circ\). The feeding ports of Ka elements are transferred to coaxial connectors, which is also beneficial to connection with TR modules. By removing a rectangular zone of copper on Ka elements, the channel isolation is better than \(-30\) dB and \(-18\) dB in Ku and Ka band, which support the simultaneous working situation for two bands. A prototype of Ku/Ka antenna array is also fabricated and measured. The measured active VSWR, channel isolation and radiating patterns are in good agreement with the simulated results, which further demonstrates its good radiating characteristics and promising application in dual-band phased-array systems.
2. Design and simulated results
The configuration of the Ku/Ka dual-band antenna is shown in Fig. 1. As shown in Fig. 1(a), the Ku elements are metallic BTAs, and the Ka elements are PCB-based tapered slot antennas. The Ku elements are periodically arranged in the gap of Ka elements, and one Ku element corresponds to four Ka elements. Due to the rectangular arrangement, the element spacing of Ku and Ka elements is 10.4*10.4 mm and 5.2*5.2 mm, respectively.
As shown in Fig. 1(b) and (c), the Ku element consists of a metallic tapered-slot radiating structure (two radiating fins), a feeding pin, and a reflecting plate ground. The detailed parameters shown in the figure are listed in the explanatory text. Recently, arrays of tapered-slot end-fire radiators show good wide band and wide-angle scanning performance. However, traditional Vivaldi antennas or antipodal Vivaldi antennas are not suitable in our case. For Vivaldi antennas, the height of elements is over 16.5 mm (~0.99 \(\lambda\)min) to achieve 4 GHz bandwidth, which will cause high cross-polarization level when scanned in the D-plane [28]. Moreover, the high profile of Ku elements will lead to serious occlusion to Ka elements, which will strongly deteriorate the Ka radiating patterns. For the antipodal Vivaldi antenna, although the profile can be reduced, balanced excitation through external baluns are required, which makes it difficult to integrate the antenna with other microwave modules. Hence, the BTA elements are the best choice for the dual-band antenna array. With the tapered slot, the antenna can match well with the free space, which will bring good radiation performance and wide bandwidth. For the BTAs, the frequency bandwidth is determined by the height (H1) and width (W1) of the radiating structure. However, the width is limited by the distance between the Ka elements, and the height need to be reduced because of its negative influence on Ka radiation pattern. In this case, the width W1 is properly increased in the upper part to broaden the bandwidth. Besides, two squared slots are designed on the edge of the tapered-slots, which will increase the equivalent path length of radiating current and decrease the height (H1) of the radiating structure effectively. With the optimization above, the Ku antenna can achieve the bandwidth of 4 GHz (relative bandwidth of 25%) and beam scanning coverage of \(\pm 60^\circ\).
As we see in Fig. 1(d), the unbalanced feeding pin shows a quite simple structure and is easy to be integrated in a coaxial cable, making it more flexible to connect the antenna with microwave modules and assemble in a phased-array radar system. Moreover, the metallic antenna structure and feeding pin shows high power capacity and good manufacturability, which is also an advantage in high-power systems.
As shown in Fig. 1(c), the Ka element is a slot antenna based on printed circuit board (PCB) and fed by strip lines. The whole thickness of the PCB is 0.608 mm, with two pieces of Rogers CLTE-XT substrates (0.254 mm thick, \(\varepsilon\)r=2.94) and one piece of Rogers FR-27 prepreg (0.1 mm thick, \(\varepsilon\)r=2.7). Other detailed parameters shown in the figure are listed in the explanatory text. As a typical tapered slot antenna, it has been demonstrated that the Ka elements have a wide band width and show good radiating characteristics [29]. In our case, the occlusion of the higher metallic Ku radiating structure may deteriorate the Ka patterns. As mentioned above, two squared slots are designed on the edge of the tapered-slots to decrease the height of the radiating structure. With this optimization, the Ka antenna can achieve the operating bandwidth of 4 GHz (relative bandwidth of 11.4%) and beam scanning coverage of \(\pm 30^\circ\). Moreover, the feeding strip lines of Ka elements offer a flexible feeding method, which makes it easy to integrate in a whole system.
Figure 2(a) and (b) show the simulated active VSWR results of Ku and Ka elements at different scanning angles. The active S-parameter is calculated by the following equation [30]:
\[\begin{equation*} \textit{ActiveS}(\textit{port } i)=\sum\nolimits_{j=1}^n S_{ij}\frac{a_j}{a_i} \tag{1} \end{equation*}\] |
Where Sij is the transmission coefficient from Port j to Port i, aj is the complex excitation of Port j, and ai is the complex excitation of Port i.
The Ku and Ka antennas operate on the band of 14 to 18 GHz and 33 to 37 GHz. As we know, the spacing between adjacent array elements is a crucial parameter for the design of an antenna array, and it directly determines the beam scanning coverage [31], [32]. In our case, the element spacing for Ku antenna is 10.4*10.4 mm, and the scanning angle can achieve 60\(^\circ\) below 15 GHz, 45\(^\circ\) below 16 GHz, and 30\(^\circ\) below 18 GHz without the appearance of grating lobes. For Ka antennas, the element spacing is 5.2*5.2 mm, and the scanning angle can achieve 30\(^\circ\) between 33 and 37 GHz.
As shown in Fig. 2, the simulated active VSWR is less than 2.8 (\(\leq 30^\circ\)), less than 2.6 (\(\leq 45^\circ\)), and less than 2.0 (\(\leq 60^\circ\)) for Ku elements, and the active VSWR is less than 2.0 for all scanning angles for Ka elements. Therefore, both Ku and Ka antenna can achieve an operating frequency bandwidth of 4 GHz. What’s more, the low VSWR indicates good radiating efficiency for both Ku and Ka antennas in the operating bandwidth.
It is worth noting that, the channel isolation between the adjacent Ku and Ka elements is also quite important for the whole system. For a phased-array radar, the dual-band antenna arrays are often required to work simultaneously at two frequency bands. Therefore, poor channel isolation will deteriorate the radiation patterns and even cause damages to the system.
In this case, the polarization of Ku and Ka antenna is designed to be perpendicular to each other, which can increase the channel isolation between two antennas. What’s more important, a rectangular zone of copper on PCB of Ka elements is removed (shown in Fig. 1(c)), which can reduce the induced current on the surface and suppress the mutual coupling effect effectively.
To study the mutual coupling effect of Ku and Ka antennas, the simulated current distribution is shown in Fig. 3.
Fig. 3 (a) (b) (c) Current distribution of Ku and Ka elements when different ports are excited. (d) Measured and simulated channel isolation of the adjacent Ku and Ka elements in two frequency bands. |
As shown in Fig. 3(a) and (b), when Ku elements are excited, the radiating current of Ku element exists on the surface of the middle slot, and very little induced current exists on the surface of Ka elements. As shown in Fig. 3(c), when Ka elements are excited, no radiating current is induced on the surface of Ku elements. The simulated results demonstrate that both Ku and Ka antenna can work independently, and the mutual coupling between them has been suppressed effectively. Figure 3(d) shows the simulated and measured channel isolation of Ku and the adjacent Ka element in two bands. We can see that both the simulated and measured channel isolation is better than \(-30\) dB in Ku band and better than \(-18\) dB in Ka band, which can support the simultaneous working situation of two bands.
Figure 4(a) and (b) show the simulated radiating pattern in an infinite periodical array at different frequencies of one Ku element and four Ka elements. Since the radiating environment is different for any of the four Ka elements due to the higher Ku radiating structure, it is more reasonable to evaluate the pattern by four Ka elements. It can be observed that the main beams of radiation pattern for Ku and Ka elements are at broadside direction without unwanted splitting or tilting. For the radiating pattern of Ka antenna, the beam width of E-plane is smaller than that of H-plane, and the gain reduction at 30\(^\circ\) is larger. This can be attributed to the Ku element’s occlusion on the radiation pattern of Ka elements. By adjusting the terminal width and height of Ku elements, the Ku bandwidth and the Ka radiating pattern can be balanced, which need to be considered in other specific circumstance.
3. Experimental results and discussion
The proposed Ku/Ka dual-band antenna array is fabricated and the prototype is shown in Fig. 5. The antenna array contains 144 Ku elements and 256 Ka elements, and the array size can be adjusted if it is necessary for measurements. The feeding pins of Ku elements are integrated in a coaxial cable, and the feeding strip lines of Ka elements are transferred to coaxial connectors, as shown in the bottom view. Owing to the convenient feeding ports, each element can be excited independently, which is essential for phased array systems. In our case, the active VSWR and channel isolation are measured with the antenna array. Meanwhile, the radiating patterns at different scanning angles are also measured and analyzed by applying different phase in the exciting ports.
The active VSWR results of Ku and Ka elements are measured, as shown in Figure 6. For Ku elements, the exciting port placed at the center of the array is selected, and other ports of Ku and Ka elements are loaded by matching loads. In the measuring process, the vector reflection coefficient of the port and the vector transmission coefficients of the port to all the other ports are measured. The active VSWR results are calculated according to Eq. (1). The active VSWR results of Ka elements are measured and calculated in the same way. For Ku elements, the measured active VSWR is less than 2.4 (\(\leq 30^\circ\)), less than 2.5 (\(\leq 45^\circ\)), and less than 3.0 (\(\leq 60^\circ\)), which is in good agreement with the simulated results. For Ka elements, the measured active VSWR is less than 2.8 for all scanning situation. Meanwhile, the channel isolation of the adjacent Ku and Ka elements in two frequency bands is measured, as shown in Fig. 3(d). Both the simulated and measured channel isolation is better than \(-30\) dB in Ku band and better than \(-18\) dB in Ka band, which can support the simultaneous working situation of Ku and Ka antennas. The measured results can demonstrate that the proposed Ku/Ka dual-band antenna can achieve a relative bandwidth of 25% and beam scanning coverage of \(\pm 60^\circ\) in Ku band and achieve a relative bandwidth of 11.4% and beam scanning coverage of \(\pm 30^\circ\) in Ka band. Besides, the Ku and Ka antenna can work simultaneously, which can meet the requirements of advanced dual-band phased-array systems.
To further investigate the radiating characteristics of the proposed Ku/Ka dual-band antenna array, the radiating patterns at different scanning angles in Ku and Ka bands are measured.
Figure 7(a) and (b) show the measured normalized radiating patterns of Ku antenna in different beam directions. The prototype is measured in an anechoic chamber, and a separated power divider network and electronical phase shifters are used to form radiating patterns in different directions. It is found that, the Ku antenna array can achieve \(\pm 60^\circ\) beam coverage in both E and H plane, and the normal gains of 14 GHz, 16 GHz and 18 Ghz are 28.69 dB, 29.84 dB and 30.96 dB, which agrees well with the simulated results. In each scanning direction, the main lobe of radiating pattern is smooth and symmetric without splitting or tilting, and no grating lobes can be observed. The simulated and measured normalized gain reduction in different scanning directions is summarized in Table I. At the same scanning angle, the gain is reducing with the frequency increasing due to the narrower beam width at high frequencies. At the same frequency, the gain reduction is roughly equal in E- and H-plane, and agrees well with the simulated results. The measured radiating pattern and gain reduction demonstrate good radiating characteristics of the Ku antenna array in both E-plane and H-plane.
Figure 8(a) and (b) show the measured normalized radiating patterns of Ka antenna in different beam directions. We can find that the Ka antenna array can achieve \(\pm 30^\circ\) beam coverage in both E and H planes, and the normal gain is 26.10 dB, 26.81 dB, and 27.43 dB for 33 GHz, 35 GHz and 37 GHz. The simulated and measured normalized gain reduction in different scanning directions is summarized in Table II. For Ka elements, the gain reduction is increasing with the frequency increasing, which is similar with Ku elements. However, the measured gain reduction in E plane at 30\(^\circ\) are larger than the simulated results. On one hand, the occlusion effect of Ku elements causes narrower Ka E-plane patterns, which has been revealed in simulation radiating patterns for Ka elements. On the other hand, Ka elements are fabricated in a small size and require higher manufacturing precision. Therefore, the fabricating error and assembly error will cause more deterioration on Ka elements, such as the active VSWR and radiating patterns. Hence, the measured and simulated results for Ka elements show more difference. In general, the beam-scanning coverage can achieve 30\(^\circ\) in both E-plane and H-plane, and better electrical characteristics can be obtained by higher manufacturing precision.
4. Conclusion
In this paper, a wide-band and wide-angle scanning Ku/Ka dual-band phased-array antenna is proposed. The dual-band antenna array consists of Ku-band metallic BTA elements and Ka-band PCB-based slot antenna elements. The Ku elements are optimized to achieve the operating bandwidth of 4 GHz and the scanning coverage of \(\pm 60^\circ\), and two square slots on the tapered edge are designed to reduce the profile. Meanwhile, the feeding pin of Ku elements is integrated in the coaxial cable, making it easy and flexible to connect to TR modules. The Ka elements are optimized to achieve the operating bandwidth of 4 GHz. Thanks to the low-profile Ku elements, the beam-scanning coverage of Ka elements achieves \(\pm 30^\circ\). The feeding ports of Ka elements are transferred to coaxial connectors, which is also beneficial to connection with TR modules. By removing a rectangular zone of copper on Ka elements, the channel isolation is better than \(-30\) dB and \(-18\) dB in Ku and Ka band, which allows the simultaneous working situation for two bands. A prototype of Ku/Ka antenna array is also fabricated and measured. The measured active VSWR, channel isolation and radiating patterns are in good agreement with the simulated results, which further demonstrates its good antenna characteristics and promising application in dual-band phased-array radar systems.
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Authors
Peng Lu
The 14th Research Institute of China Electronics Technology Group Corporation
Hao Zhou
The 14th Research Institute of China Electronics Technology Group Corporation
Zhihao Xu
The 14th Research Institute of China Electronics Technology Group Corporation
Tian Li
Southeast China Institute of Electronic Technology